AD8333 datasheet(21/28 Pages) AD

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AD8333

Rev. A | Page 21 of 28

As previously noted, a typical CW signal has a large dc and

very low frequency component compared to its desired low CW

Doppler baseband frequency, and another unwanted component

at the 2 × LO. The dc component flows through the gain resistors

R1x, while the 2 × LO flows through the capacitors C1x. The

smaller desired CW Doppler baseband signal is in the frequency

range of 1 kHz to 50 kHz.

Because the output current of the AD8333 contains the baseband

frequency, a dc component, and the 2 × LO frequency voltages,

the desired small amplitude baseband signal must be extracted

after a series of filters. These are shown in Figure 55 as LPFn,

HPFn, and gain stages.

Before establishing the value of CLPF1, the resistor RLPF1 is

selected based on the peak operating current and the linear

range of the op amp. Because the peak current for each AD8333

is 6.6 mA and there are eight channels to be summed, the total

peak current required is 52.8 mA. Approximately half of this

current is dc and the other half at a frequency of 2 × LO. Therefore,

about 26.4 mA flows through the resistor while the remaining

26.4 mA flows through the capacitor. Resistor R1 was selected

as 100 Ω and, after filtering, generates a peak dc and very low

frequency voltage of 2.64 V at the AD8021 output. For power

supplies of ±5 V, 100 Ω is a good choice for R1.

However, because the CW signal needs to be amplified as

much as possible, and the noise degradation of the signal path

minimized, the value of R1 should be as large as possible. A

larger supply helps in this regard, and the only factor limiting

the largest supply voltage is the required power.

For a ±10 V supply on the AD8021, R1 can be increased to

301 Ω and realize the same headroom as with a ±5 V supply. If a

higher value of R1 is used, C1 must be adjusted accordingly (in

this example 1/3 the value of the original value) to maintain the

desired LPF roll-off. The principal advantage of a higher supply

is greater dynamic range, and the trade-off is power consumption.

The user must weigh the trade-offs associated with the supply

voltage, R1, C1, and the following circuitry. A suggested design

sequence is:

• Select a low noise, high speed op amp. The spectral density

noise (en) should be <2nV/√Hz and the 3 dB BW ≥ 3 × the

expected maximum 2 × LO frequency.

• Divide the maximum linear output current by 6.6 mA to

determine the maximum number of AD8333 channels that

can be summed.

• Select the largest value of R1 that permits the output voltage

swing within the power supply rails.

• Calculate the value of C1 to implement the LPF corner that

allows the CW Doppler signal to pass with maximum

attenuation of the 2 × LO signal.

The filter LPF1 establishes the upper frequency limit of the

baseband frequency and is selected well below the 2 × LO

frequency, typically 100 kHz or less, or, as an example, 88 kHz

as shown in Figure 55.

A useful equation for calculating C1 is

LPF

(1)

As previously mentioned, the AD8333 output current contains a

dc current component. This dc component is converted to a

large dc voltage by the AD8021 LPF. Capacitor C2 filters this dc

component and, with R2 + R3, establishes a high-pass filter with

a low frequency cutoff of about 100 Hz. Capacitor C3 is much

smaller than C2 and, consequently, can be neglected. C2 can be

calculated by

)

(

HPF

(2)

To achieve maximum attenuation of the 2 × LO frequency, a

second low-pass filter, LPF2, is established using the parallel

combination of R2 and R3, and C3. Its −3 dB frequency is simply

() 3

LPF

(3)

In the example shown in Figure 55, fLPF2 = 81 kHz.

Finally, the feedback resistor of the AD797 must be calculated.

This is a function of the input current (number of channels)

and the supply voltage.

The second-order summing amplifier requires a very low noise

op amp, such as the AD797, with 0.9 nV/√Hz, because the

amplifier gain is determined by Feedback Resistor R4 divided

by the parallel combination of the LPF2 resistors seen looking

back toward the AD8021s. Referring to Figure 55, the AD797

inband (100 Hz to 88 kHz) gain is expressed as

[

])

(

)

(

R2B

R3A

R2A

(4)

The AD797 noise gain can increase to unacceptable levels,

because the denominator of the gain equation is the parallel

resistance of all the R2 + R3 resistors in the AD8021 outputs.

For example, for a 64-channel beamformer, the resistance seen

looking back toward the AD8021s is about 1.4 kΩ/8 = 175 Ω.

For this reason, the value of (R2x + R3x) should be as large as

possible to minimize the noise gain of the AD797. (Note that

this is the case for the AD8021 stages because they look back

into the high impedance current sources of the AD8333s.)

Due to these considerations, it is advantageous to increase the

gain of the AD8021s as much as possible, because the value of

(R2x + R3x) can be increased proportionally. Resistors (R2x +

R3x) convert the CW voltages to currents that are summed at

the inverting inputs of the AD797 op amp and amplified and

converted to voltages by R4.

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