ADP3810/ADP3811

–8–

REV. 0

charge current levels can be obtained by either reducing the

increasing R3 is lower efficiency due to the larger voltage drop

(but higher efficiency) as discussed below.

The internal band gap reference is not only used internally for

the voltage and current loops, but it is also available externally if

an accurate voltage is needed. The reference employs a pnp

output transistor for low dropout operation. Figure 3 shows a

typical graph of dropout voltage versus load current. The refer-

ence is guaranteed to source 5 mA with a dropout voltage of

400 mV or less. The 0.1

µF capacitor on the reference pin is in-

tegral in the compensation of the reference and is therefore re-

quired for stable operation. If desired, a larger value of capacitance

can also be used for the application, but a smaller value should

pin. Additional reference performance graphs are shown in Fig-

ures 2 through 6.

Output Stage

The output stage performs two important functions. It is a

buffer for the compensation node, and as such, it has a high im-

pedance input. It is also a GM stage. The OUT pin is a current

output to enable the direct drive of an optocoupler for isolated

applications. The gain from the COMP node to the OUT pin is

approximately 5 mA/V. With a load resistor of 1 k

Ω, the voltage

gain is equal to five as specified in the data sheet. A different

× (5 mA/V). Figures

20 and 21 show how the gain varies from part to part and versus

the supply voltage, respectively. The guaranteed output current

is 5 mA, which is much more than the typical 1 mA to 2 mA re-

quired in most applications.

Current Loop Accuracy Considerations

The accuracy of the current loop is dependent on several factors

tio of the internal 80 k

Ω compared to the external 20 kΩ resis-

–300 mV is guaranteed to be within 15 mV of this value. This

assumes an exact 20 k

Ω resistor for R3. Any errors in this resis-

tor will result in further errors in the charge current value. For

example, a 5% error in resistor value will add a 5% error to the

tor. Thus, 1% or better resistors are recommended.

Ω illustrates its impact on the accuracy of the

± 5 mV to

–300 mV

± 15 mV. This results in a charge current range from

250 mA

± 50 mA to 3 A ± 150 mA, as opposed to a charge cur-

rent range of 100 mA

± 20 mA to 1.2 A ± 60 mA for R

0.25

Ω. Thus, not only is the minimum current changed, but

the absolute variation around the set point is increased (although

the percentage variation is the same).

Voltage Loop Accuracy Considerations

The accuracy of the voltage loop is dependent on the offset of

GM2, the accuracy of the reference voltage, the bias current of

GM2 through R1 and R2, and the ratio of R1/R2. For the de-

manding application of charging LiIon batteries, the accuracy of

the ADP3810 is specified with respect to the final battery volt-

age. This is tested in a full feedback loop so that the single ac-

curacy specification given in the specification table accounts for

all of the errors mentioned above. For the ADP3811, the resis-

tors are external, so the final voltage accuracy needs to be deter-

mined by the designer. Certainly, the tolerance of the resistors

has a large impact on the final voltage accuracy, and 1% or bet-

ter is recommended.

Supply Range

The supply range is specified from 2.7 V to 16 V. However, a

final battery voltage option for the ADP3810 is 16.8 V. The

16.8 V is divided down by the thin film resistors to 2.0 V inter-

nally. Thus, the input to GM2 never sees much more than 2.0 V,

fixed to 2.7 V and the ADP3810 will still control the charging of

a 16.8 V battery stack. The ADP3811, with external resistors,

can charge batteries to voltages well in excess of its supply volt-

cannot be supplied directly from the battery as it is in Figure 1.

Alternative circuits must be employed as will be discussed later.

Decoupling capacitors should be located close to the supply pin.

The actual value of the capacitors depends on the application,

but at the very least a 0.1

µF capacitor should be used.

OFF-LINE, ISOLATED, FLYBACK BATTERY CHARGER

The ADP3810 and ADP3811 are ideal for use in isolated charg-

ers. Because the output stage can directly drive an optocoupler,

feedback of the control signal across an isolation barrier is a

simple task. Figure 23 shows a complete flyback battery charger

with isolation provided by the flyback transformer and the

optocoupler. The essential operation of the circuit is not much

different from the simplified circuit described in Figure 1. The

GM1 loop controls the charge current, and the GM2 loop con-

trols the final battery voltage. The dc-dc converter block is

comprised of a primary side PWM circuit and flyback trans-

former, and the control signal passes through the optocoupler.

The circuit in Figure 23 incorporates all of the features neces-

sary to assure long battery life with rapid charging capability.

By using the ADP3810 for charging LiIon batteries, or the

ADP3811 for NiCad and NiMH batteries, component count is

minimized, reducing system cost and complexity. With the cir-

cuit as presented or with its many possible variations, designers

no longer need to compromise charging performance and bat-

tery life to achieve a cost effective system.

Primary Side Considerations

A typical current-mode flyback PWM controller was chosen for

the primary control circuit for several reasons. First and most

importantly, it is capable of operating from very small duty

cycles to near the maximum designed duty cycle. This makes it

a good choice for a wide input ac supply voltage variation re-

quirement, which is usually between 70 V–270 V ac for world

wide applications. Add to that the additional requirement of

0% to 100% current control, and the PWM duty cycle must

have a wide range. This charger achieves these ranges while

maintaining stable feedback loops.

The detailed operation and design of the primary side PWM is

widely described in the technical literature and is not detailed

here. However, the following explanation should make clear the

reasons for the primary side component choices. The PWM fre-

quency is set to around 100 kHz as a reasonable compromise