OPA643 datasheet(9/16 Pages) TI1

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OPA686

transimpedance amplifier. One transimpedance design ex-

ample is shown on the front page of the data sheet. Designs

that require high bandwidth from a large area (high capaci-

tance) detector with relatively low transimpedance gain will

benefit from the low input voltage noise offered by the

OPA686. This input voltage noise will be peaked up over

frequency at the output by the diode source capacitance, and

can, in many cases, become the limiting factor to input

sensitivity. The key elements of the design are the expected

diode capacitance (CD) with the reverse bias voltage (–VB)

applied, the desired transimpedance gain, RF, and the GBP

of the OPA686 (1600MHz). Figure 3 shows a design using

a 50pF source capacitance diode and a 10k

Ω transimpedance

gain. With these three variables set (and including the

parasitic input capacitance for the OPA686 added to CD), the

feedback capacitor value (CF) may be set to control the

frequency response.

Where:

IEQ = Equivalent input noise current if the output noise is

bandlimited to F < 1/(2

πR

FCD)

IN = Input current noise for the op amp inverting input

EN = Input voltage noise for the op amp

CD = Diode capacitance

= Bandlimiting frequency in Hz (usually a post filter

prior to further signal processing)

Evaluating this expression up to the feedback pole frequency

at 15.5MHz for the circuit of Figure 3, gives an equivalent

input noise current of 6.4pA/

√Hz. This is much higher than

the 1.8pA/

√Hz for just the op amp itself. This result is being

dominated by the last term in the equivalent input noise

expression. It is essential in this case to use a low voltage

noise op amp. For example, if a slightly higher input noise

voltage, but otherwise identical op amp were used instead of

the OPA686 in this application (say 2.0nV/

√Hz ), the total

input-referred current noise would increase to 9.5pA/

√Hz.

LOW GAIN COMPENSATION FOR IMPROVED SFDR

Where a low gain is desired, and inverting operation is

acceptable, a new external compensation technique may be

used to retain the full slew rate and noise benefits of the

OPA686 while giving increased loop gain and the associ-

ated improvement in distortion offered by the decompen-

sated architecture. This technique shapes the loop gain for

good stability while giving an easily controlled second-

order low pass frequency response. Considering only the

noise gain (non-inverting signal gain) for the circuit of

Figure 4, the low frequency noise gain, (NG1) will be set by

the resistor ratios while the high frequency noise gain (NG2)

will be set by the capacitor ratios. The capacitor values set

both the transition frequencies and the high frequency noise

gain. If this noise gain, determined by NG2 = 1+CS/CF, is set

to a value greater than the recommended minimum stable

gain for the op amp and the noise gain pole, set by 1/RFCF,

is placed correctly, a very well controlled, 2nd-order low

pass frequency response will result.

FIGURE 3. Wideband, Low Noise, Transimpendance

Amplifier.

To achieve a maximally flat 2nd-order Butterworth fre-

quency response, the feedback pole should be set to:

1/(2

πR

FCF) = √(GBP/(4πRFCD))

Adding the common-mode and differential mode input ca-

pacitance (1.0 + 2.0)pF to the 50pF diode source capacitance

of Figure 3, and targeting a 10k

Ω transimpedance gain using

the 1600MHz GBP for the OPA686, will require a feedback

pole set to 15.5MHz. This will require a total feedback

capacitance of 1.0pF. Typical surface-mount resistors have

a parasitic capacitance of 0.2pF, leaving the required 0.8pF

value shown in Figure 3 to get the required feedback pole.

This will give an approximate –3dB bandwidth equal to:

f–3dB = √(GBP/2πRFCD)Hz

The example of Figure 3 will give approximately 23MHz

flat bandwidth using the 0.8pF feedback compensation.

If the total output noise is bandlimited to a frequency less

than the feedback pole frequency, a very simple expression

for the equivalent input noise current can be derived as:

FIGURE 4. Broadband Low Gain Inverting External Com-

pensation.

500

Ω

27pF

OPA686

+5V

–5V

2.9pF

250

Ω

EQ =

2 + 4kT









N 2πCD F

()2

10k

Ω

Supply Decoupling

Not Shown

50pF

OPA686

+5V

–5V

–V

O = ID RF

0.8pF

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