OPA643 datasheet(10/16 Pages) TI1

10 / 16 page

OPA686

–3

–6

–9

–12

–15

–18

Frequency (MHz)

100

500

170MHz

To choose the values for both CS and CF, two parameters and

only three equations need to be solved. The first parameter

is the target high frequency noise gain NG2, which should be

greater than the minimum stable gain for the OPA686. Here,

a target NG2 of 10.5 will be used. The second parameter is

the desired low frequency signal gain, which also sets the

low frequency noise gain NG1. To simplify this discussion,

we will target a maximally flat second-order low pass

Butterworth frequency response (Q = 0.707). The signal

gain of –2 shown in Figure 4 will set the low frequency noise

gain to NG1 = 1 + RF/RG (= 3 in this example). Then, using

only these two gains and the GBP for the OPA686

(1600MHz), the key frequency in the compensation can be

determined as:

Physically, this Z0 (10.6MHz for the values shown above) is

set by 1/(2

π • R

F(CF + CS)) and is the frequency at which the

rising portion of the noise gain would intersect unity gain if

projected back to 0dB gain. The actual zero in the noise gain

occurs at NG1 • Z0 and the pole in the noise gain occurs at

NG2 • Z0. Since GBP is expressed in Hz, multiply Z0 by 2π

and use this to get CF by solving:

Finally, since CS and CF set the high frequency noise gain,

determine CS by [Using NG2 = 10.5]:

The resulting closed-loop bandwidth will be approximately

equal to:

For the values shown in Figure 4, the f–3dB will be approxi-

mately 130MHz. This is less than that predicted by simply

dividing the GBP product by NG1. The compensation

network controls the bandwidth to a lower value while

providing the full slew rate at the output and an excep-

tional distortion performance due to increased loop gain at

frequencies below NG1 • Z0. The capacitor values shown

in Figure 4 are calculated for NG1 = 3 and NG2 = 10.5 with

no adjustment for parasitics.

Figure 5 shows the measured frequency response for the

circuit of Figure 4. This shows the expected gain of

–2 (6dB) with exceptional flatness through 70MHz and a

–3dB bandwidth of 170MHz. Measured distortion into a

100

Ω load shows > 5dB improvement through 20MHz over

the performance shown in the Typical Performance Curves.

Into a 500

Ω load, the 5MHz, 2Vp-p, 2nd harmonic improves

from –90dBc to –96dBc.

O =

GBP

1–









–1 – 2













FIGURE 5. G = –2 Frequency Response with External

Compensation.

LOW NOISE FIGURE, HIGH DYNAMIC RANGE “IF”

AMPLIFIER

The low input noise voltage of the OPA686 and its high

two-tone intercept can be used to good advantage as a fixed

gain IF amplifier. While input noise figures in the 10dB

range (for a matched 50

Ω input) are easily achieved with

just OPA686 alone, Figure 6 shows a technique which

reduces the noise figure even further while providing a

broadband, low gain IF amplifier stage using the OPA686.

FIGURE 6. Low Noise Figure IF Amplifier.

Bringing the signal in through a step-up transformer to the

inverting input gain resistor has several advantages for the

OPA686. First, grounding the non-inverting input elimi-

nates the contribution of the non-inverting input current

noise to the output noise. Secondly, the non-inverting

input voltage noise of the op amp is actually attenuated if

reflected to the input side of RG. Using the 1:2 (turns ratio)

step up transformer reflects the 50

Ω source impedance at

F =

2π •R

F Z O NG

S =

2 –1

() C

–

3dB ≅

O GBP

(= 2.86pF)

(= 27.2pF)

(= 130MHz)

OPA686

+5V

–5V

200

Ω

2pF

20pF

Ω Source

Ω Load

1:2

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