OPA830IDBVT datasheet(23/35 Pages) TI

23 / 35 page

OPA830

SBOS263F − AUGUST 2004 − REVISED AUGUST 2008

www.ti.com

MACROMODEL AND APPLICATIONS

SUPPORT

Computer simulation of circuit performance using SPICE

is often a quick way to analyze the performance of the

OPA830 and its circuit designs. This is particularly true for

video

and

amplifier

circuits

where

parasitic

capacitance and inductance can play a major role on

circuit performance. A SPICE model for the OPA830 is

available through the TI web page (www.ti.com). The

applications department is also available for design

assistance. These models predict typical small signal AC,

transient steps, DC performance, and noise under a wide

variety of operating conditions. The models include the

noise terms found in the electrical specifications of the

data sheet. These models do not attempt to distinguish

between the package types in their small-signal AC

performance.

OPERATING SUGGESTIONS

OPTIMIZING RESISTOR VALUES

Since the OPA830 is a unity-gain stable, voltage-feedback

op amp, a wide range of resistor values may be used for

the feedback and gain setting resistors. The primary limits

on these values are set by dynamic range (noise and

distortion) and parasitic capacitance considerations. For a

noninverting unity-gain follower application, the feedback

connection should be made with a direct short.

Below 200

Ω, the feedback network will present additional

output loading which can degrade the harmonic distortion

performance of the OPA830. Above 1k

Ω, the typical

parasitic capacitance (approximately 0.2pF) across the

feedback resistor may cause unintentional band limiting in

the amplifier response.

A good rule of thumb is to target the parallel combination

of RF and RG (see Figure 3) to be less than about 400Ω.

The combined impedance RF || RG interacts with the

inverting input capacitance, placing an additional pole in

the feedback network, and thus a zero in the forward

response. Assuming a 2pF total parasitic on the inverting

node, holding RF || RG < 400Ω will keep this pole above

200MHz. By itself, this constraint implies that the feedback

resistor RF can increase to several kΩ at high gains. This

is acceptable as long as the pole formed by RF and any

parasitic capacitance appearing in parallel is kept out of

the frequency range of interest.

In the inverting configuration, an additional design

consideration must be noted. RG becomes the input

resistor and therefore the load impedance to the driving

source. If impedance matching is desired, RG may be set

equal to the required termination value. However, at low

inverting gains, the resultant feedback resistor value can

present a significant load to the amplifier output. For

example, an inverting gain of 2 with a 50

Ω input matching

resistor (= RG) would require a 100Ω feedback resistor,

which would contribute to output loading in parallel with the

external load. In such a case, it would be preferable to

increase both the RF and RG values, and then achieve the

input matching impedance with a third resistor to ground

(see Figure 9). The total input impedance becomes the

parallel combination of RG and the additional shunt

resistor.

BANDWIDTH vs GAIN:

NONINVERTING OPERATION

Voltage-feedback op amps exhibit decreasing closed-loop

bandwidth as the signal gain is increased. In theory, this

relationship is described by the Gain Bandwidth Product

(GBP) shown in the specifications. Ideally, dividing GBP

by the noninverting signal gain (also called the Noise Gain,

or NG) will predict the closed-loop bandwidth. In practice,

this only holds true when the phase margin approaches

°, as it does in high-gain configurations. At low gains

(increased feedback factors), most amplifiers will exhibit a

more complex response with lower phase margin. The

OPA830 is compensated to give a slightly peaked

response in a noninverting gain of 2 (see Figure 3). This

results in a typical gain of +2 bandwidth of 110MHz, far

exceeding that predicted by dividing the 110MHz GBP by

2. Increasing the gain will cause the phase margin to

approach 90

° and the bandwidth to more closely approach

the predicted value of (GBP/NG). At a gain of +10, the

11MHz bandwidth shown in the Electrical Characteristics

agrees with that predicted using the simple formula and

the typical GBP of 110MHz.

Frequency response in a gain of +2 may be modified to

achieve exceptional flatness simply by increasing the

noise gain to 3. One way to do this, without affecting the +2

signal gain, is to add an 2.55k

Ω resistor across the two

inputs, as shown in Figure 7. A similar technique may be

used to reduce peaking in unity-gain (voltage follower)

applications. For example, by using a 750

Ω feedback

resistor along with a 750

Ω resistor across the two op amp

inputs, the voltage follower response will be similar to the

gain of +2 response of Figure 2. Further reducing the value

of the resistor across the op amp inputs will further dampen

the frequency response due to increased noise gain. The

OPA830 exhibits minimal bandwidth reduction going to

single-supply (+5V) operation as compared with

±5V. This

minimal reduction is because the internal bias control

circuitry retains nearly constant quiescent current as the

total supply voltage between the supply pins is changed.

INVERTING AMPLIFIER OPERATION

All of the familiar op amp application circuits are available

with the OPA830 to the designer. See Figure 9 for a typical

inverting configuration where the I/O impedances and

signal gain from Figure 1 are retained in an inverting circuit

configuration. Inverting operation is one of the more

common requirements and offers several performance

benefits. It also allows the input to be biased at VS/2

without any headroom issues. The output voltage can be

independently moved to be within the output voltage range

with coupling capacitors, or bias adjustment resistors.

OPA830IDBVT Datasheet(PDF) 23 Page - Texas Instruments

OPA830IDBVT Datasheet(HTML) 23 Page - Texas Instruments

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