AD8309 datasheet(14/20 Pages) AD

14 / 20 page

REV. B

AD8309

–14–

load current, which may be large, the value of R2 should take

this into account.

The four pins labeled PADL tie down directly to the metallic

lead frame, and are thus connected to the back of the chip. The

process on which the AD8309 is fabricated uses a bonded-wafer

technique to provide a silicon-on-insulator isolation, and there is

no junction or other dc path from the back side to the circuitry

on the surface. These paddle pins must be connected directly to

the ground plane using the shortest possible lead lengths to

minimize inductance.

Basic Connections

Figure 30 shows the connections required for most applications.

The inputs are ac-coupled by C1 and C2, which normally

should have the same value, say, CO. The coupling time con-

stant is ROCO /2, where RO = RS + RIN, thus forming a high pass

corner with a 3 dB attenuation at fHP = 1/(

π R

T CC ). In high-

frequency applications, fHP should be chosen as large as pos-

sible, to minimize the coupling of unwanted signals. On the

other hand, in low frequency applications, a simple RC network

forming a low-pass filter should be added at the input for the

same reason.

VLOG

VPS2

PADL

LMHI

LMLO

PADL

FLTR

LMDR

COM2

VPS1

PADL

INHI

INLO

PADL

COM1

ENBL

AD8309

0.1 F

RLOAD

RLIM

RSSI

LMHI

LMLO

0.1 F

ENABLE

SEE TEXT FOR MORE

ABOUT DECOUPLING

SIGNAL

INPUTS

52.3

4.7nH

FOR BROADBAND 50

TERMINATION TO 1GHz

NC = NO CONNECT

Figure 30. Basic Connections

Where it is necessary to terminate the source at a low imped-

ance, the resistor RT should be added, with allowance for the

shunting effect of the 1 k

Ω input resistance (R

IN) of the AD8309.

For example, to terminate a 50

Ω source, a 52.3 Ω resistor

should be used for signal frequencies up to about 50 MHz. The

termination means may be placed either at the input or at the

log amp side of the coupling capacitors. In the former case

smaller capacitors can be used for a given frequency range; in

the latter case, the dc resistance is lowered directly at the log

amp inputs, which helps to keep offsets to a minimum. At

higher frequencies, the reactance of the 2.5 pF input capaci-

tance must be accounted for. A 4.7 nH inductor in series with

the 52.3

Ω termination resistor provides an essentially flat 50 Ω

input impedance to 1 GHz. An impedance-transforming net-

work is preferably used to provide a 50

Ω interface, since this

also introduces a balanced voltage gain of typically 13 dB and

the AD8309 has a very high capacity for large input voltages.

Figure 31 shows the output versus the input level, with the axis

marked in dBm (correct only when terminated in 50

Ω), for sine

inputs at 5 MHz, 50 MHz, 100 MHz and 200 MHz. Figure 32

shows the typical logarithmic linearity (law conformance) under

the same conditions.

INPUT LEVEL – dBm Re 50

2.5

–100

2.0

1.5

1.0

0.5

–80

–60

–40

–20

100MHz

50MHz

200MHz

5MHz

Figure 31. RSSI Output vs. Input Level at TA = +25°C, for

Frequencies of 5 MHz, 50 MHz, 100 MHz and 200 MHz

INPUT LEVEL – dBm Re 50

–90

–1

–2

–3

–4

–5

–80 –70 –60 –50 –40 –30 –20 –10

DYNAMIC RANGE

5MHz

50MHz

100MHz

200MHz

3dB

103

102

1dB

50MHz

5MHz

200MHz

100MHz

Figure 32. Log Linearity vs. Input Level at TA = +25°C, for

Frequencies of 5 MHz, 50 MHz, 100 MHz and 200 MHz

Input Matching

Where either a higher sensitivity or a better high frequency

match is required, an input matching network is valuable. Using

a flux-coupled transformer to achieve the impedance transfor-

mation also eliminates the need for coupling capacitors, lowers

any dc offset voltages generated directly at the input, and use-

fully balances the drives to INHI and INLO, permitting full

utilization of the unusually large input voltage capacity of the

AD8309.

The choice of turns ratio will depend somewhat on the fre-

quency. At frequencies below 30 MHz, the reactance of the

input capacitance is much higher than the real part of the input

impedance. In this frequency range, a turns ratio of 2:9 will

lower the effective input impedance to 50

Ω while raising the

input voltage by 13 dB. However, this does not lower the effect

of the short circuit noise voltage by the same factor, since there

will be a contribution from the input noise current. Thus, the

total noise will be reduced by a smaller factor. The intercept at

the primary input will be lowered to –120 dBV (–107 dBm).

Impedance matching and drive balancing using a flux-coupled

transformer is useful whenever broadband coupling is required.

However, this may not always be convenient. At high frequen-

cies, it will often be preferable to use a narrow-band matching

network, as shown in Figure 33, which has several advantages.

First, the same voltage gain can be achieved, providing increased

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