ADP3182 datasheet(18/20 Pages) AD

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ADP3182

Rev. 0 | Page 18 of 20

A type three compensator on the voltage feedback is adequate

for proper compensation of the output filter. Equations 20 to 22

are intended to yield an optimal starting point for the design;

some adjustments may be necessary to account for PCB and

component parasitic effects.

OUT

(17)

⎟

⎠

⎞

⎜

⎝

⎛

−

OUT

(18)

(19)

If CX is 6000 µF (five 1200 µF capacitors in parallel) with an

equivalent ESR of 3 mΩ, the equations above give the following

compensation values:

CA = 1.33 nF

RA = 6.05 kΩ

CFB = 110 pF

Using the nearest standard value for each of these components

yields CA = 1.2 nF, RA = 6.04 kΩ, and CFB = 100 pF.

INPUT CAPACITOR SELECTION AND

INPUT CURRENT di/dt

In continuous inductor current mode, the source current of the

high-side MOSFET is approximately a square wave with a duty

ratio equal to n × VOUT/VIN and an amplitude of one-nth the

maximum output current. To prevent large voltage transients, a

low ESR input capacitor, sized for the maximum rms current,

must be used. The maximum rms capacitor current is given by

0.1

−

CRMS

(20)

Note that manufacturers often base capacitor ripple current

rating on only 2,000 hours of life. Therefore, it advisable to

further derate the capacitor or to choose a capacitor rated at a

higher temperature than required. Several capacitors may be

placed in parallel to meet size or height requirements in the

design. In this example, the input capacitor bank is formed by

two 2,700 µF, 16 V aluminum electrolytic capacitors and three

4.7 µF ceramic capacitors.

To reduce the input current di/dt to a level below the recom-

mended maximum of 0.1 A/µs, an additional small inductor

(L > 370 nH @ 10 A) can be inserted between the converter and

the supply bus. That inductor also acts as a filter between the

converter and the primary power source.

INDUCTOR DCR TEMPERATURE CORRECTION

With the inductor’s DCR being used as the sense element and

copper wire being the source of the DCR, one needs to

compensate for temperature changes in the inductor’s winding

if a highly accurate safety current limit setpoint is desired.

Fortunately, copper has a well-known temperature coefficient

(TC) of 0.39%/°C.

If RCS is designed to have an opposite and equal percentage of

change in resistance to that of the wire, it cancels the tempera-

ture variation of the inductor’s DCR. Due to the nonlinear

nature of NTC thermistors, resistors RCS1 and RCS2 are needed.

See Figure 10 for instructions on how to linearize the NTC and

produce the desired temperature tracking.

ADP3182

CSCOMP

CSSUM

CSREF

RTH

RCS1

RCS2

CCS1

CCS2

RPH1

RPH2

RPH3

SWITCH

NODES

VOUT

SENSE

PLACE AS CLOSE AS POSSIBLE

TO NEAREST INDUCTOR

OR LOW-SIDE MOSFET

KEEP THIS PATH

AS SHORT AS POSSIBLE

AND WELL AWAY FROM

SWITCH NODE LINES

Figure 10. Temperature Compensation Circuit Values

The following procedures and expressions yield values to use

for RCS1, RCS2, and RTH (the thermistor value at 25°C) for a given

RCS value.

Select an NTC based on type and value. Because we do not

have a value yet, start with a thermistor with a value close

to RCS. The NTC should also have an initial tolerance of

better than 5%.

Based on the type of NTC, find its relative resistance value

at two temperatures. The temperatures that work well are

50°C and 90°C. These resistance values are called A

(RTH(50°C)/RTH(25°C)) and B (RTH(90°C)/RTH(25°C)). Note that the

NTC’s relative value is always 1 at 25°C.

Find the relative value of RCS required for each of these

temperatures. This is based on the percentage of change

needed, which in this example is initially 0.39%/°C. These

are called r1 (1/(1 + TC × (T1 − 25))) and r2 (1/(1 + TC ×

(T2 − 25))), where TC = 0.0039 for copper. T1 = 50°C and

T2 = 90°C are chosen. From this, one can calculate that r1 =

0.9112 and r2 = 0.7978.

Compute the relative values for RCS1, RCS2, and RTH using

(

)

()

(

)

()

(

)

CS2

−

(21)

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