Copyright © 2017, Texas Instruments Incorporated

Bias Current

Cancellation

and R-noise

filtering

2.4-Mhz

Butterworth

+

1-MHz

Low Pass

PD

+

±

U1 OPA838

VM1

V1 3 V

LM7805 ± 230 mV

+

+

R1 20

R2 73.2

33

OPA838

www.ti.com

SBOS867 – AUGUST 2017

Product Folder Links: OPA838

Submit Documentation Feedback

Copyright © 2017, Texas Instruments Incorporated

9.1.6 Transimpedance Amplifiers

A common application for a high gain bandwidth voltage feedback op amp is to amplify a small photo-diode

current from a capacitive detector. Figure 80 shows the front page transimpedance circuit with more detail. Here,

a fixed –0.23 negative voltage generator (LM7705) is used on the negative supply to ensure the output has

adequate headroom when it is at 0 V. The transimpedance stage is designed here for a 2.4Mhz flat (Butterworth)

response while a simple RC post-filter band-limits the broadband noise and sets the overall bandwidth to 1MHz.

The requirements for a high dynamic range transimpedance (or charge) amplifier include the very low input

voltage noise intrinsic to a decompensated device like the OPA838. The noise gain over frequency for this type

of circuit starts out at unity gain then begins to peak with a single zero response due to the pole formed in the

feedback by the feedback resistor and the total capacitance on the inverting input. That noise gain response is

flattened out at higher frequencies by the feedback capacitor value to be the 1+Cs/Cf capacitor ratio. This is

normally a very high noise gain allowing the decompensated OPA838 to be applied to this application. Since the

noise gain is intentionally peaked to a high value in this application, the very low input voltage noise (1.8 nV/

√Hz)

of the OPA838 improves dynamic range.

Figure 80. 100-kΩ Wide Bandwidth Transimpedance Design

9.1.7 Design Requirements

To implement a controlled frequency response transimpedance design, set the transimpedance stage amplifier

bandwidth higher than a controlled post RC filter. This allows variation in the source capacitance and amplifier

gain bandwidth product with less overall bandwidth variation to the final output. In this example design:

•

Assume a nominal source capacitance value of 100 pF. This normally comes from the capacitance versus

reverse bias plot for the photodiode. No reverse bias is shown in Figure 81, but the current source is typically

a back biased diode with a negative supply on the anode and the cathode connected to the op amp inverting

input. In this polarity, the signal current sinks into the diode and raises the op amp output voltage above

ground.

•

For the best DC precision, add a matching resistor on the noninverting input to reduce the input bias current

large as a capacitor as required to roll off resistor noise. This capacitor has a relatively low frequency self

resonance that interacts with the input stage and might impair stability. Add a small series 20-Ω resistor from

the capacitor into the noninverting input to de-Q the resonant source impedance without adding too much

noise.

•

Set the feedback capacitor to achieve the desired frequency response shape.

•

Add a post RC filter to control the overall bandwidth to 1 MHz. In this example, a 2.2-nF capacitor allows a

low 73.2-Ω series resistor. When driving a sampling ADC (like a SAR), this combination helps reduce the