LTC7813 datasheet(22/28 Pages) AD

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LTC7878

Rev. 0

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APPLICATIONS INFORMATION

affect the output voltage regulation as long as the other

input sources can supply enough load current. The peak

inductor currents are still shared among all the buck-

boost converters by tying all the ITH pins together. In the

redundancy design, it is suggested that each LTC7878

has its own compensation network and feedback resistor

locally closed to the pin and then short the VFB pins and

ITH pins all together with PCB traces.

DESIGN EXAMPLE

As a design example for a buck-boost converter, assume

VIN(NOMINAL) = 8V ~ 42V, VOUT = 12V, IOUT = 9A, and

fSW = 250kHz. Maximum ambient temperature = 60°C.

1. Set the frequency at 250kHz by applying 1V on the

FREQ pin (See Figure 2), 10μA current flowing out of

the FREQ pin will give 1.0V across a 100k resistor to

GND.

RFREQ (in kΩ) =

1.0V

10µA

= 100kΩ

2. Determine the inductor value. The nominal input volt-

age is 24V and set inductor ripple current to 60% of DC

current (5.4A) at the nominal input voltage. This leads

to an inductor value:

= VIN − VOUT

(

)• DTS

= 24V − 12V

(

) •

0.5 •

4µs

5.4A

= 4.4µH

The peak inductor current at 24VIN is 11.7A.

To achieve high efficiency, low DCR inductors are

preferred. We choose Wurth 7443630420 inductor

(4.2μH/3.04mΩ) with 24A rated current. With 2 RC

filters and usage of ISNSD pin, we can set the peak

sensed inductor current limit to 50mV. Then the peak

inductor current is 50mV/3.04mΩ = 16.45A which is

less than the saturation current of the selected inductor.

Next the inductor current at minimum and maximum

input voltage should be examined. At VIN = 8V and

VOUT = 12V, the converter is working as a boost con-

verter. The peak inductor current with 9A load current

is 14.7A. At VIN = 42V and VOUT = 12V, the converter

is working as a buck converter. The peak inductor cur-

rent with 9A load is 13.1A. Both are less than the peak

current limit of 16.45A.

The DCR current sensing filter network should be

designed based on the Figure 8. if C1 = C2 = 47nF,

then R1 = L/(4 • DCR)/C1 = 7.35kΩ, R2 = R1 • 3 • 0.9

= 19.845kΩ. In real design, choose R1 = 7.32kΩ and

R2 = 20kΩ. The worst case minimum sensed ripple

signal for the internal current comparator happened at

boost mode with 16.67% duty cycle. In this case, the

inductor ripple current peak to peak value is 1.6A and

sensed voltage is ∆VSENSE = 4 • DCR • ∆IL(MIN) = 19mV.

The sensed voltage is higher than 15mV so the signal

to noise ratio is high enough and the jitter is minimized.

3. Set the output voltage. Select feedback resistors R1

and R2. If R1 is 10k, R2 is:

VOUT •R1

1.0

– R1

Select R2 as 110k. Both R1 and R2 should have a tol-

erance of no more than 1%.

4. Select MOSFET based on the maximum input and

output voltages. Switches A and B connected to SW1

operates under the max 42V input voltage, so the 60V

rated MOSFETs are selected. Switches C and D con-

nected to SW2 operates under the 12V output voltage,

so the 25V rated MOSFETs are selected. Similar to buck

or boost converter design, the low RDS(ON) MOSFETs

are preferred for low conduction losses. However, the

switching loss may be high for low RDS(ON) MOSFETs

due to the large gate capacitance. LTC7878 can drive

both logic level and non-logic level MOSFETs for users

to select the MOSFET widely. In the design example,

BSC034N06NS3 (non-logic level) and BSC014NE2LSI

(logic level) are used. Bootstrap diodes should be able

to handle the maximum VIN/VOUT voltage with higher

than 100mA capability. PMEG6010 Schottky diodes are

selected for the TG1/TG2 drivers.

Since the normal level MOSFETs are selected, an

input voltage UVLO may be set up on the RUN pin. A

100k/20k resistor divider from VIN to RUN pin will turn

on the converter when VIN is higher than 7.2V.

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